What is to be considered as motor current limitation

Motor control with PWM


When dimensioning the current-carrying capacity of the components, two main values ​​have to be considered:

  • Current consumption of the motor in normal operation (i.e. with the load)
  • Current consumption with blocked motor or starting current. Here the current is only limited by the ohmic resistance in the circuit, so it can quickly reach the 2-3-digit ampere range, depending on the motor.

The usual dimensioning is mainly based on the first value and plans appropriate reserves (factor 2-5 of the current for a few seconds), since the second value is usually difficult to achieve and unnecessary in practice for reasons of cost. Therefore, the problem is avoided either with a fast current limitation or with a soft start by slowly increasing the PWM. This limits the maximum current to a significantly lower value so that the components can be weaker and more cost-effective.

The voltage at the motor and thus approximately the idle speed is proportional to the duty cycle:

[math] \ displaystyle {\ text {Motor voltage} = \ text {Operating voltage} \ cdot \ mathrm {Tastverh \ ddot {a} ltnis}} [/ math]

Circuit variants

Mosfet with free-wheeling diode, 1-quadrant controller

The simplest circuit consists only of transistor T1, the motor, the freewheeling diode D1, the capacitor C1, as well as the actual PWM generation and the MOSFET driver.

During the switch-on phase of T1, the entire operating voltage is applied to the motor. The difference between the generator voltage generated by the rotating motor and the operating voltage drops across the winding resistance and winding inductance. So you are dealing with an RL series connection. Since the winding resistance is very small, the current increases approximately linearly until T1 switches off. Then D1 takes over the current flow and closes the circuit until T1 takes over the current again (or the current has decayed = intermittent operation). Although no more energy is supplied from the outside, the motor is still supplied by the energy stored in the winding. The current now drops linearly again until T1 switches on again and supplies energy again.

The amplitude of the current through the diode D1 is exactly the same as the current through T1, but depending on the duty cycle, the rms value is smaller than that through T1 (with over 50% duty cycle), the same (with 50% duty cycle) or even greater ( with a duty cycle less than 50%). Therefore, the diode must be dimensioned as strong as the transistor (if possible use a fast switching diode (e.g. Schottky), no slow rectifier diodes such as 1N400x). The capacitor C1 is necessary to smooth the current pulsed by the PWM, because otherwise a voltage drop or voltage peaks would occur in the supply lines due to the steep edges (it is best to use 2 or more capacitors of different capacities and types, usually a film capacitor with a few Microfarads in parallel with an aluminum electrolytic capacitor).

The transistor, the diode and the capacitor should be placed as close together as possible (shortest connections). If the diode is attached to the motor, the lead emits massive interference. The lines should best be twisted.

Synchronous rectification, 2-quadrant controller

To reduce the losses in the freewheeling diode, you can replace it with a second Mosfet, which is always switched on when the other one is switched off.

Since this mosfet, unlike a diode, conducts current in both directions, it is possible to short-circuit the motor and thus brake the motor. In general, one can say: If the mean voltage generated by the PWM is greater than the generator voltage of the motor, it will be accelerated. If the voltage is lower, the motor is braked. The motor is short-circuited by T2, so that the generator voltage initially reduces the current in the motor winding and then builds it up again in the opposite direction. When T2 is switched off and T1 switched on, this current flows back via T1 into C1 and thus the voltage source. The energy is not destroyed when braking, but converted back into electrical energy. One should consider this when braking a large mass, because the voltage source must be able to absorb the energy. Should the voltage source z. B. consist of a transformer with a rectifier, it cannot absorb the energy but only C1, which means that the operating voltage increases until one of the transistors is destroyed. To prevent this, an overvoltage limiter in the form of a braking resistor must be provided (braking chopper). In simple cases, a power Zener diode or a suitable replica of a Zener diode and power transistor is sufficient. If the voltage source exists z. B. from a battery then this takes the energy and is charged again when braking. However, this only works if the pulse duty factor is not 0%, i.e. switch T2 and T1 alternately so that not all of the energy in the motor winding and T2 is burned. You should therefore avoid changing the duty cycle quickly in one of the two directions, as this leads to a high current.

The control circuit must also prevent T1 and T2 from becoming conductive at the same time, because this would lead to a short circuit in the operating voltage. A fully assembled Mosfet driver prevents this.

H-bridge, 4-quadrant adjuster

The H-bridge or the 4-quadrant controller is an extension of the 2-quadrant controller with a second half-bridge. In addition to accelerating and braking the motor, this also enables the direction of rotation to be reversed. There are several control methods for this

  1. The most efficient is to operate one half as with the 2-quadrant controller and to connect the second motor connection to the operating voltage with the other. For the other direction of rotation, you simply change the halves, i.e. connect the other connection to the operating voltage and use the other half as a 2-quadrant controller.
  2. The other process alternately controls T1 and T4 or T2 and T3, i.e. always applies a voltage to the motor. If the duty cycle is 50%, an average current of 0A flows because the motor receives a positive and a negative voltage for half of the time, i.e. the motor is at a standstill. Depending on whether you choose the duty cycle above or below, you determine the direction of rotation. This method is simpler in terms of control, but also generates switching losses in the transistors and losses in the motor when the vehicle is at a standstill.

Summary of properties

1 quadrant plate

  • Speed ​​specification in one direction

2 quadrant disks

  • Speed ​​specification in one direction
  • Active braking possible
  • Energy can be fed back from the motor into the power supply (recuperation)
  • slightly lower losses than 1 quadrant regulator

4 quadrant disks

  • Speed ​​specification in two directions
  • Active braking possible in two directions
  • Energy can be fed back from the motor into the power supply (recuperation)
  • about twice as large losses as 2 quadrant regulators

Example circuits

1-quadrant controller with discrete MOSFET driver

Simple 1-quadrant controller

This circuit is suitable for motors up to about 35V and 10A continuous current. Q1 and Q2 together with their wiring serve as level converters from 3.3 or 5V digital signals to 12V for the Mosfet Gate. The circuit works inverting, so the Mosfet switches on when there is a low at the input. If Q1 blocks, Q2 is turned on via R1 and supplies about 11-11.5V to the gate of the Mosfet. R2 limits the current. The Mosfet is switched off via path Q1 and D1. At the same time, the base voltage is removed from Q2 so that it blocks. D2 between the base and collector of Q1 prevents it from reaching saturation, so that it locks almost instantaneously as soon as the input changes to low. From a few amps, the freewheeling diode D3 needs a small heat sink, as does Q3. If the operating voltage V + of the motor is around 10-16V, then this voltage can also be used for the Mosfet control. Otherwise, a separate voltage of around 12-15V should be used for this.

1-quadrant controller with discrete high-side MOSFET driver

Simple 1-quadrant controller with P-channel Mosfet

This circuit is suitable for motors with an operating voltage of around 15-40V and a continuous current of up to 10A. Circuits with P-channel mosfets should be avoided if possible, since N-channel mosfets, due to their principle, have values ​​that are 3 times better than P-channel mosfets. However, sometimes an output referenced to ground is required, then this circuit is the right one. The problem with the high-side control is that you have to somehow limit the gate-source voltage to a maximum of 20V, which is not so easy with an additional auxiliary voltage of 12V as in the previous circuit. Therefore a different approach is being taken here. Q1 and R2 form a constant current source. Because there is a fixed logic voltage at the base, Q1 controls so strongly that it reduces the base voltage itself due to the voltage drop across R2. The current flowing is (logic level at the input - 0.6V base emitter voltage) / R2. A good value for the current is around 10-15mA. When dimensioning you should also consider the power loss: 40V * 15mA = 0.6W. This is far too much for a transistor in the TO92 package. Because Q1 works in linear mode, it does not reach saturation and the diode can be dispensed with, as in the previous circuit. A base resistor is also unnecessary or even out of place here. Since it is now known that in the controlled state of Q1 at 5V logic voltage (5V-0.6V) / 330Ω = 13.3mA, you can use it to calculate the voltage drop across R1, or from the desired voltage R1: U (R1) = 13.3mA * 1kOhm = 13.3V. When switched on, the voltage at the collector of Q1 is 13.3V more negative than V + and that is independent of the operating voltage! Since this voltage is buffered by the transistors Q2 and Q3, which are connected as emitter followers, and is also applied to the gate of the MOSFET, it receives around 13V gate voltage when switched on. This works well, because for most mosfets you should use a value between 10 and 15V. The voltage divider from R1 and R2 also results in a lower limit of the operating voltage so that the circuit works properly: To get the 13.3V via R2, or the 4.4V via R1, at least 17.7V for V + is necessary.

2-quadrant controller with half-bridge MOSFET driver

Simple 2-quadrant controller

This circuit is suitable for motors up to about 35V and 20A continuous current. In order to minimize the effort, a finished half-bridge driver IR2184 is used for the half-bridge. This has an integrated dead time of 500ns between switching the Mosfets, so that switching on both Mosfets at the same time is impossible. Since the Mosfet is connected to V +, this circuit also works inverting. The reason why the motor is not connected to GND is as follows: To control the highside mosfet Q2, a voltage of around 10V more than the operating voltage V + is necessary. This is generated from C2 and D1 via the bootstrap circuit. If Q1 is fully controlled, C2 is charged via D1. If Q1 then switches off and Q2 on, its control voltage is taken from C2. Since Q2 now connects VS to V +, the potential at C2 also increases. At the VB pin there are now about V + + 12V-0.7V (12V operating voltage-flow voltage from D1). Due to leakage currents, however, C2 discharges within a few milliseconds. The maximum on-time of Q2 is therefore limited. 100% duty cycle would therefore not be possible. To avoid this problem, the motor is switched to V +, so that Q1 has to be switched on for 100% duty cycle, which is not a problem, because its driver is supplied directly to the 12V. If, on the other hand, Q1 is permanently off, i.e. Q2 on, the motor brakes or stands still so that no current flows through the MOSFETs. So it doesn't matter if Q2 switches off again after a short time. The only thing that is not possible is permanent braking of the motor using Q2, but in practice this is only required in the rarest of cases. The inverted PWM signal with logic levels is fed in via the IN pin. Both Mosfets can be switched off together via the SD \ Pin. This enables the motor to coast down without braking (freewheeling). From around 5 amps, the mosfets need a small heat sink.

Choice of PWM frequency

When choosing the PWM frequency, you have to consider several factors and make a compromise:

  • The motor inductance L smoothes the current, the winding resistance R leads to a drop in the current, which results in the electrical time constant of the motor [math] \ displaystyle {t = \ frac {L} {R}} [/ math]. With many motors this is around 1ms. In the case of high-quality motors, this information should be found in the data sheet. In order to keep the current ripple low, i.e. to keep the torque constant, the period duration of the PWM should not exceed this time. This is particularly important in 2 or 4-square-meter operation, because there the current can also change its direction, which leads to deceleration, and thus to significant jerking and vibrations of the motor and unnecessary losses.
  • Frequencies between 100Hz and 10kHz generate audible whistling noises in the motor
  • With increasing frequency, the switching losses in the transistors and the losses in the armature winding and in its core increase.
  • Simple DC motors (brush motors) with built-in interference suppression capacitors can only be operated with relatively low PWM frequencies of approx. 30-200 Hz. At higher frequencies there are strong switching losses due to the reloading of the interference suppression capacitors, as you can see in this article.

Ideal frequency

The ideal PWM frequency from an electrical point of view is therefore usually 1-2 kHz. However, this is exactly the area where the hearing is most sensitive. So if the whistle of the engine doesn't bother you, this is the ideal area. Since the lower frequency is limited by the electrical time constant of the motor, you can only move upwards. A compromise is therefore the 5-15kHz range in which the hearing is significantly less sensitive and the losses are still kept within limits. If you use the simple circuit with the freewheeling diode and do not value smooth running, or if you have a high mass on the motor so that it is sluggish, then you can reduce the PWM frequency to 200Hz as an alternative to the> 5kHz to make the noise more bearable. However, the efficiency is then reduced due to the high current ripple. In the case of high-current applications, switching losses are an additional problem.

See also